0 TO 5V OUTPUT ANALOG HALL SENSOR FOR FOOT CONTROLLER

What a “0–5 V” module really outputs

A pedal sensor sold as 0–5 V output is usually a ratiometric linear Hall IC, sometimes with an op-amp. The Honeywell SS495A runs from 4.5 V to 10.5 V, idles at 2.50 V ±0.075 V, gives 3.125 mV/G ±0.125 mV/G over a ±600 G minimum linear range, and spans 0.2 V to (Vs − 0.2 V) typically, 0.4 V to (Vs − 0.4 V) guaranteed.

So about 8 % of the rail at each end is unusable, and past the linear range the output clamps: an over-travelling pedal reads “fully down” while the mechanism is elsewhere.

The ADC reference must be the same node as the sensor supply

Ratiometric means quiescent voltage, sensitivity and clamp levels all scale with the supply: V_out = 0.5·V_s + (3.125 mV/G · V_s/5 V)·B. Divide by V_s and the supply cancels. It only works if the ADC converts against the same voltage the sensor runs from.

On an ATmega32 the REFS bits in ADMUX select AREF, AVCC, or the internal 2.56 V bandgap. AVCC must stay within ±0.3 V of VCC and wants a low-pass filter. Tie AREF to the sensor’s 5 V rail through that filter and a 5 % rail change moves signal and reference together: the code stays put.

Pick the internal 2.56 V reference and cancellation is gone twice over. A 2.5 V quiescent output converts to about 999 counts of 1023, so the top of the travel is clipped, and each percent of rail drift becomes a percent of measured travel. The converter cannot measure above its reference either, so a 5 V sensor on a 3.3 V MCU needs a divider.

Counts, linearity and temperature drift

Ten bits over 5 V is 4.88 mV per LSB — about 1.56 G per count at 3.125 mV/G. A ±250 G stroke covers 1.56 V, roughly 320 counts, or 0.3 % of travel per step: the converter, not the sensor, sets resolution. Linearity is −1.0 % of span typical, −1.5 % maximum, a few counts of bow that a calibration table absorbs. Finer steps need the Allegro A1324 at 5.0 mV/G (0.98 G per LSB) or a 12-bit ADC.

Temperature moves both ends. The SS495A specifies ±0.06 %/°C null drift and −0.01 %/°C to +0.05 %/°C sensitivity drift; the A1324 quotes −0.03 %/°C. Honeywell notes that low-cost magnets have a negative remanence coefficient which the sensor’s positive sensitivity coefficient only partly compensates. Null drift is what the driver feels: a pedal that no longer reads zero. Fix it with a start-up zero: sample the released position, compare with the previous run — but require the reading to sit near the bottom for a second first, so a resting foot cannot be learned as zero.

Magnet geometry sets the travel mapping

The element responds to the field component perpendicular to the branded face, and field versus distance is not linear: flat and saturated close in, falling steeply — close to inverse-cube — once the gap is large against the magnet. Slide a magnet past a fixed sensor, or rotate a diametrically magnetised cylinder on the shaft, where the sensed component follows roughly a sine and is near-linear only near the zero crossing.

You are measuring flux density, not position: map one to the other with a table, two points if the mechanism is good, eight to sixteen if not. The element has no hysteresis; anything you see comes from the spring, the pivot or a loose magnet.

Filtering versus latency

The sensor is fast (3 µs response, 7 mV peak-to-peak noise on the A1324 with a 0.1 µF bypass), so the ADC and filter set the bandwidth. A 10 kΩ/100 nF low-pass has a 159 Hz corner and about 1 ms of group delay. The ATmega32 wants a 50–200 kHz ADC clock and 13 clocks per conversion (25 for the first after enabling or changing channel): at 125 kHz a sample takes about 104 µs, and eight averaged add 0.8 ms. That lands near 2 ms, comfortable in a 10 ms control period and hopeless behind a 10 Hz filter, which alone costs 16 ms.

Put the analogue pole just above the mechanical bandwidth you need, sample at twice that, and take a median of three before averaging.

Fail-safe when a wire opens

The two open-circuit faults push the signal in opposite directions; both look like real positions. Open the sensor ground and its output stage cannot sink; the pin is dragged up through the ADC’s protection structures toward the supply, so the controller reads full throttle. Open the supply and the output falls to zero: it reads released. No software separates “full pedal” from “broken ground” on one channel.

• A 1–10 kΩ series resistor at the ADC pin, limiting clamp current when the sensor is unpowered and the MCU is not.
• A pull-down at the pin, so a disconnected signal reads zero instead of floating.
• A window check: the SS495A cannot reach either rail, so codes near 0 or 1023 are wiring faults.
• A rate limit — the stroke cannot be covered in one 10 ms sample.
• Two tracks with opposite slopes, cross-checked, as production pedal assemblies use.

The figures above come from the cited datasheets; the arithmetic follows. Gap, table points, filter corner and fault thresholds belong to a specific mechanism and board: establish them on your hardware, at both stops and with the harness unplugged.

References

• Honeywell Sensing, SS490 Series Miniature Ratiometric Linear Hall-effect Sensor ICs, datasheet, SS495A.
• Allegro MicroSystems, A1324/A1325/A1326 Low Noise Linear Hall Effect Sensor ICs, datasheet.
• Microchip Technology (Atmel), ATmega32/ATmega32L, datasheet 2503, ADC and ADC Noise Canceler chapters.